The present invention relates to a skew estimator for estimating a skew between a first signal of a first data path and a second signal of a second data path in a coherent receiver. Further, the present invention relates to a skew compensator for compensating the estimated skew and to a coherent receiver, in particular a coherent optical receiver, including a skew estimator and a skew compensator.
An important goal of long-haul optical fiber systems is to transmit the highest data throughput over the longest distance without signal regeneration in optical domain. Because of given constraints on the bandwidth imposed by optical amplifiers and ultimately by the fiber itself, it may be important to maximize spectral efficiency. Most systems use binary modulation formats, such as on-off keying encoding one bit per symbol.
According to references [1]-[6], advanced modulation formats in combination with coherent receivers enable high capacity and spectral efficiency. Polarization multiplexing, quadrature amplitude modulation and coherent detection may provide a winning combination for high-capacity optical transmission systems since they allow information encoding in all available degrees of freedom.
Further, commercial devices using QAM constellation are available in 40 and 100 Gb/s optical systems.
In this regard, FIG. 15 shows a schematic block diagram of a coherent optical receiver 1500. The coherent optical receiver 1500 has a receive (Rx) analog part 1501 and a receive (Rx) digital part 1503.
The Rx analog part 1501 has a local oscillator (LO) 1505 and a 90° hybrid 1507 having two poles. The hybrid 1507 receives the optical signal. Four optical front ends (OFE) 1509, 1511, 1513, and 1515 are coupled to the hybrid 1507. Each OFE block 1509-1515 is coupled to one automatic gain control (AGC) block 1517, 1519, 1521, and 1523. Further, each AGC block 1517-1523 is coupled towards an analog-digital converter (ADC) 1525, 1527, 1529, and 1531. In detail:
Since the digital signal is mapped into both polarizations, the 90° hybrid 1507 is used to mix the input optical signal with a local oscillator (LO) signal of the LO 1505 that results in four output signals, namely two signals per polarization. The optical OFEs 1509-1515 are configured to convert the respective electrical signal into an optical signal. The respective OFE 1509-1515 may comprise a photo diode and a transimpedance amplifier (TIA). Because the signal power may vary over time, the AGC blocks 1517-1525 may compensate for signal power variations (see reference [7]). The four AGC blocks 1517-1525 may also be an internal part of the OFE blocks 1509-1515.
Due to realization complexity, a pair of AGC blocks may be controlled by one control signal. For example, the pair of AGC blocks 1517, 1519 may be controlled by the control signal VXAGC for X polarization. Further, the pair of AGC blocks 1521, 1523 may be controlled by the control signal VYAGC for Y polarization. Further, the four AGC blocks 1517-1523 may be controlled by four independent control voltages or control signals.
The signals output by the AGC blocks 1517-1523 may be quantised by the ADCs 1525-1531. The four ADCs 1525-1531 may output an X-polarized in-phase signal (XI), an X-polarized quadrature-phase signal (XQ), a Y-polarized in-phase signal (YI) and a Y-polarized quadrature-phase signal (YQ).
Further, the four quantised digital data streams XI, XQ, YI and YQ are further processed in a digital signalling processing (DSP) block 1533 of the Rx digital part 1503. The DSP 1533 may comprise a software part 1535 and a hardware part 1537. The hardware part 1537 may be fast compared to the slow software part 1535. The DSP block 1533 may be configured to compensate for chromatic dispersion (CD), polarization mode dispersion (PMD), polarization rotation, non-linear effects, LO noise, LO frequency offsets and the like. Moreover, an estimation of slow processes, like LO frequency offsets or CD, may be done in the software part 1535 of the DSP block 1533.
Further, FIG. 16 shows a schematic block diagram of basic DSP blocks 1600. The DSP 1600 has a software part 1601 and a hardware part 1603. The hardware part 1603 has an offset and gain adjustment (AGC) block 1605.
Coupled to the AGC block 1605, there are two compensation blocks 1607 and 1609, namely a chromatic dispersion (CD) block for X polarization 1607 and a CD compensation block 1609 for Y compensation.
Further, the hardware block 1603 comprises a frequency recovery block 1611 and a polarization mode dispersion (PMD) and chromatic dispersion (CD) compensation and depolarization block 1613 coupled to the recovery block 1611. The PMD/CD compensation and depolarization block 1613 may comprise a finite impulse response (FIR) filter.
Moreover, a timing estimation block 1615 receives the outputs of the CD compensation block 1609 and the PMD/CD compensation and depolarization block 1613 for providing a timing information towards a VCC 1617.
After the block 1613, a carrier recovery block 1619 is coupled to a decoding detection block 1621.
Further, between the data paths providing the input signals X, XQ, Y and YQ, there are four ADCs 1623, 1625, 1627 and 1629 coupled. In detail:
After offset and gain correction by block 1605, the four signals are equalized for chromatic dispersion in frequency domain using the two fast Fourier transformation (FFT) blocks 1607 and 1609. The frequency offset may be removed in the frequency recovery block 1611. Polarization tracking, PMD compensation and residual CD compensation may be done in time domain using FIR filters 1613, exemplarily arranged in a butterfly structure.
The carrier recovery block 1619 is configured to provide residual frequency offset and carrier phase recovery. When differential decoding is applied at the transmitter side (not shown), a differential decoder may be used in the decoding and frame detection block 1621.
Further, CD may efficiently be compensated in the FFT blocks 1607 and 1609. The compensation CD function may be
                                          CD                          -              1                                ⁡                      (            DL            )                          =                  exp          (                                    -                                                j                  ⁡                                      (                                                                  2                        ⁢                        π                        ⁢                                                                                                  ⁢                                                  nf                          s                                                                    N                                        )                                                  2                                      ⁢                                                            λ                  0                  2                                ⁢                DL                                            4                ⁢                π                ⁢                                                                  ⁢                c                                              )                                    (        1        )            where λ0 is the signal wavelength, fs is the sampling frequency, N is the FFT size, c is the speed of light, n is the tap number, L is fiber length, and D is the dispersion coefficient.
Due to complexity reasons, only one FFT block 1701 using complex input may be applied to each polarization, as exemplarily shown in FIG. 17. The inverse FFT (IFFT) 1703 may be identical to the FFT 1701 although real and imaginary parts are swapped at input and output.
Between the FFT block 1701 and the IFFT block 1703, an inverse chromatic dispersion (CD−1) block 1705 is coupled.
The four data paths, as exemplarily shown in FIG. 16, may have different length or delay. As a consequence, different arriving instances cause penalties, in particular significant penalties depending on the actual conditions of the channel and the amount of data delays. For example, in 112 G QPSK transmission systems with a symbol length of about 36 ps, the penalties due to I/Q skew are shown in FIG. 18. In FIG. 18, the x-axis shows the I/Q skew and the y-axis shows the required OSNR at BER of 0.001. It may be noted that a skew of 5 ps may result in 1 dB OSNR penalty. This skew value may be expected in 112 G coherent receivers. The skew may come from different transfer functions of the 90° hybrid, the OFE, the AGC, the ADC and connections between them. Furthermore, the skew between two polarizations may not be so difficult because it may be manifest as an additional differential group delay (DGD). This skew may be compensated by the use of the FIR filter. However, in this case, the DGD operating range may be decreased by the skew value. As a result, it may be desired to compensate also for the X/Y skew.
On the other hand, skew effects clock recovery performance when all four data paths are used for timing extraction.
It may be more difficult when residual dispersion should be compensated after the FFT block in the FIR block. Results for 112 G QAM with RD of −340 ps/nm are presented in FIG. 19. In particular, FIG. 19a shows the signal constellation with CD without skew and FIG. 19b shows the same for a skew of 8 ps. Further, it may be noted that a Y polarization may have more problems than an X polarization even after 50 FIR filter updates.
Regarding skew compensation, the path delay offsets between XI, XQ, YI and YQ data may be measured by applying an identical optical signal, in particular a single polarization, to all photo detectors during factory calibration. This may be done by turning off the LO and increasing the power of the signal. A common, directly detected signal may be incident on all photo detectors. Then, data blocks may be transferred to a personal computer (PC), either from before a variable FIFO buffer or after fixed filters, with all filters providing a single impulse response.
Then, the data may be interpolated and then cross-correlated between the four data paths. The relative peaks may be used to determine the relative time offsets. In particular, FIFO may be compensated for the minimum skew of the sampling period.